Interstage coupling network having improved phase response



June 15, 1954 w. E. BRADLEY INTERSTAGE COUPLING NETWORK HAVING IMPROVEDPHASE RESPONSE 2 Sheets-Sheet l Filed Aug. l1, 1950 Afro/QM?? June 15,'1954 w, E, BRADLEY 2,681,391 INTERSTAGE COUPLING NETWORK HAVING IMPROVEDPHASE RESPONSE Filed Aug. 11, 1950 2 Sheets-Sheet 2 Patented June 15,1954 INTERSTAGE COUPLING NETWORK HAVING i IMPROVED PHASE RESPONSEWilliam E. Bradley, Newtown,

Pa., assignor to Philco Corporation, Philadelphia, Pa., a corporation ofPennsylvania ApplicationAugust 11, 1950, Serial No. 178,784

` 5 Claims. 1

The invention herein described and claimed 'relates to electrical signaltransfer networks, and in particular to such networks which areparticularly adapted for use in the intermediate 'frequency ampliercircuits of television receivers.

In conventional television transmission, the several channels assignedfor use by different transmitting stations are located closely adjacenteach other in the frequency spectrum. This imposes stringentrequirements upon the design of television receiving equipment, andparticularly on the design of intermediate frequency amplifiers employedin such equipment, if undesirable interference between signals inadjacent channels is to be avoided. More particularly, the location ofthe sound carrier of one channel is ordinarily located very close to oneextremity of the adjacent picture channel. This requires that theintermediate frequency amplifier `of a television receiver be designednot only to transmit effectively the frequency components within thedesired picture channel, but also to provide maximum rejection of theundesired adjacent sound carrier frequency. To this end, it has beencustomary in the past to provide, in television intermediate frequencyamplier circuits, a rejection or trap .circuit to provide high`attenuation of this undesired sound carrier frequency. Such circuitsare customarily resonant circuits, of one form or another, havingextremely high `sensitivity, by reason of which they tend undesirably tomodif-y the frequency response characteristic of the intermediatefrequency amplifier Within the desired picture channel. `Moreparticularly, they have a tendency substantially to reduce the fre-Yquency response at the extremity of the channel which is adjacent tothe undesired sound carrier frequency. `It has been customary in theprior .art to compensate for this undesired modification in the picturechannel response by the inclusion Iof circuits to provide an additionalpole or resonance in the `vicinity of this extremity of the picturechannel to boost the response in that region. Unfortunately thisexpedient has had the undesirable effect of adversely modifying thephase response characteristic of the intermediate frequency amplifierwithin the desired picture channel so .as to `render it non-linear.While various expediente have been resorted `to in the `.past toovercome this difculty, none of Vthem have proved. entirelysatisfactory.

Accordingly it is the primary object of my ,invention to provide anelectrical signal transfer network for :providing a desired signaltransfer characteristic throughout a predetermined iirst frequency band,for providing high attenuation of signals in a second closely adjacentband, and for providing a desirably linear phase response characteristicthroughout said first frequency band.

More particularly, it is an object of my invention to provide `animproved television intermediate frequency amplifier having apredetermined desired and preferably'uniform amplitude versus frequencyresponse characteristic `for signal frequency components Within a givenchannel, having a substantially linear phase characteristic forfrequency components within said channel, and adapted effectively toreject a sound carrier frequency located in an adjacent channel.

Brieiiy, these objectives are achieved in accordance with the presentinvention by the inclusion in the signal transfer network of circuitswhich are eiective to produce a pole in its transfer impedancecharacteristic which lies in the band between the extremity of thedesired passband and the frequency to which the adjacent channelrejection or trap circuit is tuned. Such a circuit differs from those ofthe prior art, as above discussed, in which circuits were included forproducing an additional pole within the desired passband and in thevicinity of the extremity closest to the frequency to which therejection or trap circuit was tuned. Such an arrangement, it has beenfound, .is effective to compensate for the undesired modification of thefrequency response characteristic within the -desired passband which isproduced by the inclusion of the adjacent channel rejection circuit,While at the same time maintaining a suitably linear phase responsecharacteristic Within the desired Dassband.

Before proceeding to the detailed description of the various embodimentsof my invention, it will be -conducive to a .better understandingthereof, to review some of the fundamental concepts relating thereto.

To begin with, it Will be recalled that the transfer impedance of anetwork isthe ratio of the output voltage of the network produced inresponse to a predetermined input current vsupplied thereto. This isalgebraically represented by the expression Where Zr is theaforementioned transfer impedance,` E@ is the `output voltage and Ii isthe input current of the network.

methods of circuit analysis and ,is hereinafter y carried out, by way ofexample, for several actual networks.

It is then further well known that such an expression for'Z'rl1 canalways be put into the form (-oiN-ozN-Ms) (2) 7\p1)()\ }\1127)(}`)113)where A is a complex quantity of the form e-I-ye; and K is a constantnot involving A.

Incidentally y' and w are quantities well known in the art, i being thesquare root of l and w being the product of frequency and 27T.

The numerator of Equation 2 and, with it, the transfer impedance ZTevidently goes to zero Whenever A equals A A0 A03, and so forth.Similarly the denominator of Equation 2 goes to zero and the transferimpedance goes to infinity when A equals Apl, Apg, Aps, etc. A pole issaid to exist, in the transfer impedance characteristic, at eachfrequency for which the latter condition obtains, namely for vwhich Aequals Apl, APE, Ap, etc. At these frequencies, the transfer impedancepasses through a maximum. On the other hand, a zero is said to exist ateach frequency for which the former-condition obtains, namely for whichA=A A02, A03, etc. At zero frequencies, the transfer impedance passesthrough a minimum.

Since any given network gives rise t0 only one expression ofthe form ofEquation 2, such a network has a transfer impedance characterized by adistinctive pole and zero pattern. From this pattern, both the amplitudeand phase response of the network may be computed by simple graphicalmethods which are in current use in a Wide segment of the network designfield. Thus, a given pole and Zero pattern is characteristic of a, givenamplitude and phase response of a network. Conversely, a network may befully described, so far as its significant operational characteristics,namely, phase and amplitude response are concerned, by specifying thepole and z ero pattern of its transfer impedance characteristic.

As hereinbefore indicated, the objectives of my invention are realizedby a network whose transfer impedance characteristic has a pole and zeropattern distinguished by the presence of a pole within the range offrequencies intermediate the end of the ampliner passband and theadjacent channel trap circuit.

For a better understanding of the reasons for this choice of polepattern and of the novelty residing therein, reference may now be had tothe subsequent discussionY in conjunction with the Vaccompanyingdrawings, wherein:

Figure 1 is illustrative of the theoretical ccnsiderations underlyingthe novel features of my network;

Figure 2 shows the pole and Zero pattern of a network constructed inaccordance with the invention as well as its Vdeparture from the priorart;

Figure 3 shows a basic embodiment of a network constructed in accordancewith the invention; and

Figure 4 shows a preferred embodiment of the invention.

Referring more particularly to Figure l of the drawings, there is showntherein a system of orthogonal coordinates as commonly used for thegraphical representation of the poles and zeros of electrical networks.Since, as hereinbefore described, poles and zeros are dened by certaincritical values of A in the solution of Equation 2, it suffices, for acomplete definition of each pole and zero, to plot the real andimaginary parts of the complex expression Yfor each such critical valueof A. Accordingly, the vertical axis of the coordinate system of Figure1 is calibrated in terms of the real portion of A, which is a ashereinbefore defined, increasing values of @being plotted in an upwarddirection from the origin of the coordinate system. The imaginary partof A, which is je, is plotted along the horizontal axis, increasingpositive values of y'w extending to the right of the aforementionedorigin. In this connection it is useful to note that values of A plottedin this system of coordinates have a very real meaning in terms ofthetransmission characteristics of the network with which they areassociated. Thus the position of the plotted value of A measured alongthe je axis is directly proportional to and may be interpreted in termsof the frequency at which the particular pole, or zero as the case maybe, occurs, while its position along theV c axis is proportional to themagnitude of the resistiveattenuation which the network presents tosignals to be transmitted therethrough at the particular frequencyinvolved. Thus, the further the pole, or zero, is displaced from theorigin along the a axis the higher will be the attenuation of thenetwork on signals transmitted therethrough and the lower the Q of thenetwork. Broadly speaking, then, the proximity of the pole or zero tothe origin measured along the a axis is indicative of the selectivity ofthe circuit which produces the pole or zero, increasing displacementfrom the originV indicating decreasing selectivity of the network.

Proceeding now to the practical application of these criteria it will,first of all,v be shown how pole and zero placement affects the phaseand amplitude response of electrical networks. For this illustrativepurpose a simple network consisting of an inductance and a capacitorconnected in parallel has been chosen, this particular arrangement ofcircuit elements being basic to television I.F. amplifiers.Incidentally, although such a parallel L.C. network may not be providedwith a separate resistor, some resistance will always be presenttherein, owing to the inherent resistance of the inductor and possiblyto some leakage resistance in the capacitor. This resistive componentmay for all practical purposes, be considered as a lumped resistance inparallel with the capacitor and the incluctor. Its presence gives riseto the resistive losses which lower the Qfof the network from what thesame would be in the absence of such resistance.

A parallel L.C. network as here described is generally called a singletuned circuit. As may readily be demonstrated, such a single tunedcircuit has a transfer impedance which is characterized by a single poleor zero, as the case may be, on the positive side of the y'w axis of thecoordinate system shown in Figure l. Whether this network produces apole or a zero depends upon the manner in which it is connected in thepath of signals to be transmitted therethrough. Speciiically, if theentire network is connected in 'shunt Awith this signal path then `itstransfer impedance will have a maximum, orpcle, where- .asifit is`connected in series with the signal path it will be characterized by aminimum, or zero.

This is readily apparent from elementary tuned evidencing. the presenceof a zero. rI'his again is,

of course, substantiated by mathematical analy- It will be seen, fromthe foregoing discussion, that a single tuned resonant circuit seriallyconnected in the signal path readily lends itself to `use as an adjacentchannel trap circuit, as hereinbefore defined, and such a circuit is infact so utilized in certain embodiments of my invention. Such a trapcircuit has, preferably, low resistive losses for then its amplituderesponse will be extremely low for signals of its zero frequency. Sinceit is normally dimcult to obtain `resonant circuits having such lowlosses and since such low losses with their attendant high Q and goodattenuation are essential for use with adjacent channel trap circuitsfor television I.-F. amplifiers, special precautions were taken in thisparticular case to reduce the resistive losses of the trap circuit to aminimum. The detailed manner of carrying out this neutralization of thelosses is, incidentally, well known and does not constitute a part of myinvention. Its detailed `explanation is therefore relegated to a laterpoint in this discussion. Suffice it to say, for the time being, thatthe losses of the trap circuit are extremely low with the result thatthe saine is highly selective and strongly attenuates signals at itszero frequency. As indicated, it is` precisely under these conditions oflow losses or high selectivity that the improvements effected by mynovel circuit arrangement are most conspicuous.

Considering now the phase shift to which a particular frequencycomponent will be subjected by the poles or zeros variously disposedwith respect to it of a signal transfer network traversed `by it, itwill be seen` from the diagram of Fig. l, that the phase shift to whicha frequency component corresponding to the point labeled je, on

.the horizontal axis will be subjected by a complex pole pl having thecoordinates up, and impl, is

equal to the angle 0., included between a line connecting the pole plwith iwi, and a perpendicular from the pole to the je axis. Similarlythe phase `shift to which the same frequency component will Ibesubjected by a pole p2 having the coordinates up, and jam, which isfurther displaced along the horizontal axis, is equal to the angle 0,included between a line connecting the pole p2 with im and aperpendicular from this pole to the jw axis. Thus it is seen that therate of change of phase shift with frequency due to a `given poledecreases rapidly with increasing separation of the two along thehorizontal axis. However it is also to be noted that, if the pole lieson the horizontal axis, it will have a constant 'phase-shifting effectat frequencies on thefsame horizontal axis. Further it will be notedthat zeros in the transfer impedance characteristic of the network willhave similarly varying phasei .I

shifting `effects depending `upon their locations with respect to thefrequency componentin question. However, while a pole located at a givenpoint will produce a phase shift in one sense, a zero located at thesame point will produce a `like phase shift in the opposite sense.

Next, with regard to the effect of a particular pole or zero on theamplitude of .a given frequency component, it is to be noted that, whilethis likewise diminishes with increasing separation, between the pole orzero and the frequency component, measured along the horizontal axis,ther-ate of decrease is not as rapid as it is in the case of the effectof the pole or zero on phase.

Consider now the application of these principles to the prior art and tocircuits in accordance with the present invention. In the prior art itwas considered necessary to provide a pole in the transfer impedancecharacteristic of a television intermediate frequency amplifier at apoint within the desired picture frequency band and adjacent theextremity of that band nearest to the undesired adjacent channel soundcarrier `frequency in order to compensate for the tendency of the soundcarrier rejection circuit tomodify the amplitude response of the`picture channel in that region. By reason of the location of this polewithin the band, it had a substantial effect in adversely modifying thephase response characteristic for signal frequency components withinsaid band. In accordance with the present invention, however, theadditional pole in the transfer impedance `characteristic is located notwithin the desired `picture frequency passband, but rather between theupper extremity of that band and the frequency of the adjacent channelsound carrier. rThis is clearly illustrated in Fig. 2 which shows thebasic pole and zero pattern of the transfer impedance of a televisionintermediate frequency amplifier constructed in accordance with thepresent invention. As shown in this diagram, this characteristic has apole p3 located near the lower extremity of the picture frequencypassband and a zero located at the frequency je@ corresponding to thefrequency of the adjacent sound frequency carrier to be rejected, as inthe case of prior art circuits. However, the pattern is furthercharacterized in having an additional pole p4 located in the regionbetween the upper extremity of the passband and the adjacent soundcarrier frequency.

In this respect, the pattern differs signicantiy from the pole and zeropattern of a conventional intermediate frequency amplier circuit `inaccordance with the prior art. Although this additional pole is locatedexternally of the picture frequency passband, it still has,` as aboveindicated, 'a substantial effect in compensating the reduction in theamplitude response throughout the higher frequency portion of thepicture frequency passband which the zero of the ad'- jacent channelsound rejection circuit tends to produce. However, by reason of itslocation in this position its effect on the phase characteristic withinthe picture frequency passband is substantially less than would be thecase if it were Ilocated within that passband, and, in particular,

:it does not appreciably adversely affect the linearity of the phasecharacteristic.

Thus it is seen that while the prior art teaches that the additionalpole should be located within the picture .frequency passband in orderto securethe'desired compensation of the amplitude responsecharacteristic within that band, I have taught that substantially thesame results in re- Vcated within the passband, this discrepancy may becompensated for by appropriately increasing the Q of the circuits whichproduce the pole, and hence the same overall compensating effect may beobtained as is obtained when a pole is -located internally of thepassband.

It will be understood that, in the actual design of an amplifier inaccordance with the invention, the particular pole and zero positionsselected may be governed to a large extent by ancillary considerations,provided only that the aforedescribed pole placement outside thepassband is secured.

Proceeding now to the consideration of actual circuit arrangementsembodying my inventive concept, reference may nrst be had to f Figure 3of the drawings where there is illustrated a basic form which such anembodiment may take in practice. The circuit there illustrated is thatof a television I.F. ampliiier having four stages of amplificationinterconnected by three separate coupling networks. The signal path thusformed begins at the control grid of triode Vi to which 1.-?. signalsderived from preceding stages of the receiver are applied, and leadsfrom there through triodes Il, I2 and I3, all connected in cascade, tothe output of the network whence it is supplied to whatever circuitsconventionally follow the I.F. ampliner. A parallelinductance-capacitance network, or single tuned circuit i4, shunts thesignal path between the anode of triode l0 and the grid of triode l i.Another single tuned circuit .l5 shunts the signal path intermediate theanode of triode il and the control grid of triode I2, while the thirdsingle tuned circuit IS is connected in series with the signal pathbetween the anode of triode i2 and the control grid of triode I3. lnaddition, suitable conventional anode load resistors are provided foreach triode whereby the respective anodes are connected to a commonsource of positive potential B+. Similarly, the control grid electrodesof all the triodes are connected, through suitable individual grid leakresistors, to a common source of negative grid potential C. ConventionalD.C. blocking capacitors are, of course, also provided, intermediate theanode of each triode and the control grid of the following triode, toprevent the deleterious application of B+ potential to the latter.

In this network the only elements which are frequency sensitive aresingle tuned circuits i4, It and it, the vacuum tubes being provided notonly for the purpose of amplification of the signal but also to preventinteraction between these tuned circuits which would disturb theirsimple single tuned characteristic.

As has been explained, hereinbefore, a single tuned circuit connected inshunt with the signal path has a transfer impedance characterized by asingle pole at the resonant frequency of the circuit, whereas a singletuned circuit connected in series with a signal path has a transferimpedance characterized by a single Zero, again at its resonantfrequency. Accordingly, in the embodiment under consideration, tunedcircuits I4 and I5 are tuned, respectivel to the frequencies of the twopoles which my coupling network must have to conform with my inventiveconcept, whereas tuned circuit I3 is made resonant at the zero frequencyof my network. In the practical case of an I.-F. amplifier fortelevision have a passband of 4.5 megacycles extending between thelimits of 22.1 megacycles and 26.6 megacycles and which is required toeliminate signals of the adjacent channel sound frequency of 28.1megacycles, tuned circuit Iii may be made resonant at a frequency nearthelower end of this passband and within the passband; a typical valueto which this circuit may be tuned for good results is 26 megacycles.The adjacent channel sound trap circuit, which is, in this embodiment,constituted by tuned circuit I5 in series with the signal path is, ofcourse, made resonant at the frequencyV of the adjacent channel soundcarrier, in this case 28.1 megacycles. The remaining resonant circuit,namely tuned circuit iii connected in shunt with the signal path is, inaccordance with the invention, made resonant at a frequency lyingbetween the end of the passband and the adjacent channel trap frequency,in this case between 25.6 megacycles and 28.1 megacycles. A suitablefrequency to which such a circuit may be turned, within this range, is27.6 megacycles, this being sufficiently close to the adjacent channeltrap frequency to provide all the advantages conferred by a circuitconstructed in accordance with my teachings.

It is emphasized, in this connection, that the particular values ofresonant frequencies given are simply illustrative of one particulararrangement which has beensuccessfully used, it being well understoodthat specic requirements relating to the amplitude and phase response ofthe amplifier within its passband may be met by suitable adjustment ofthe resonant frequencies of the tuned circuits as well as of theirresistive attenuation characteristics, within the limits of myteachings. Further, it will be understood that the positions of thevarious tuned circuits in the network of Fig. 3 may be interchangedwithout, in any way, deleteriously affecting the operation of thecircuit.

As has been seen, the simplied embodiment shown in Fig. 3 requires, inaddition to the three tuned circuits which produce the pole and zeropattern, four vacuum tubes to produce the necessary isolationtherebetween. Since it is often desirable to reduce the cost of suchequipment, as well as its complexity, by maintaining the number ofcircuit components at a minimum, the preferred embodiment of theinvention employs only two vacuum tubes in addition to three tunedcircuits required to produce the pole and zero pattern.

A specific circuit of this nature is illustrated in Fig. 4 to which moredetailed reference may now be had. The signal path formed by thisampliner is seen to Ycomprise an input triode Il to whose control gridelectrode I.F. input signals derived from preceding stages of thereceiver are supplied. Anode signals produced in response thereto arethen transmitted to the control grid of output triode i8 from whoseanode, in turn, corresponding I.-F. output signals are derived. Thenetwork interconnecting the anode of tube Il and the control grid oftube it comprises, in this preferred embodiment, a pair of single tunedcircuits I 8 and 2G connected in shunt with the signal path, togetherwith another single tuned circuit ZI connected in series with the signalpath. It

acs 1,391

zuillfbernoted .thatpin thisrparticularembodiment, circuits yl9cand112:are arrangedttoebe resonant :the (seriesrconnectedmsingle:tunedecircuitiszscme- :atithe` same frequency-,and fto; have :substantially hwhatmodified, Strom :its conventional form, if by aequalresistive glosses,:we may conclude?l that::their esubstitutingifor the normallyusedsinglecapac- :individual xxpoles will `be at lathe esame :locationizztorianwequivalentfseries combination of two ca- V5 S0 that l-pac'itors,` respectively designatedrbyreference nu- A (7) irmerals 122::andil 'from whose `jnnctionia resistor 1"- 2 Muis :connected finfrshuntwiththe ,signalfpatlr Substituting in Equation 6 we obtain Zte'MlCwO2o(-)\1l )2+4021(019+020) @"Ng) (X-Mi) 41(8) "lihemanfnerinrwhich thisarrangement con Observe that,;1infEquationzf8,ithe numerator;=.is"tributes to the 'satisfactory'foperationof `the cir- 15.nowalreadyfinthey;formfprescribedfzbyeEquation jcuitisgexplainedIdetailhereinafter- In ad- 2 for the determination of zero locations.Ac-

*ditiomthere "are,^ofcourse,j provided thei usual cordingly,.the:zerofofrthe entire network '-vwill "ancillaryrcircuitcomponentssuchas-"andderload be located at A2 or, in other words, at thereswresistors vconnectingeach"of-triodes"Wand I8 to ionant frequency oftuned-circuit 2| alone, which a source of positive' anodepotentialrli,ra`iD.-`C. 20 thus constitutes thetrap circuitof thenetwork. blocking capacitor 25 connected in series with the As has beenpointed out, hereinbeforelfitiis signal `path for the lpurpose 'of-preventing the desired, :.in .the case. cina televisionL-F..iampliianode potentialottriode l1 from nbeingapplied lergthatrthe zerodetermined-bythis'traprcirltothe'control grid-'of tubelf8,aswelleassuitable cuit be located as close to the fw axis as practi-"grid'biasresistorsco1ineoting eachoftriodes Il 25,:cal, sothatthesignaly.attenuationrduentoithe andwtoa source of constantnegativegridbias trapcircuit will be amaximum. Itisto achieve upotentialC- It Will-nowbeshown how-the crithe necessary. low` resistive losses that .the un-'teriaot pole analysis hereinbefore setfor-thrand usual ,arrangement `oftuned circuit f2l hereinasrestricted in'accordancewith the inventionyarebefore `outlined is provided. In'thisearrange- 'applied tothe' networkVunder considerationso as 30 ment,'.the resistor 24producesabridgeelikeac- Ito-determine all' of its `essential designrcharacter-`tion.causing. currentsA flowing oppositeldirec- `istics. tionsin thetuned'circuitito 4canc'el,softhatlfno "For-this we `rst write theequation forthe net potential is developed across the resistance,ztransfer "impedance 'ofthe network, yas hereininherent in the coils-of` the tuned circuit. This before dened in"Equation 'l `Vand Vweobtain, in 35 .results inthe desired minimizing :of `#resistiveaccordance -rwith `-w'ell*known principles of hnetlosses in the circuit`sothatritsefiectivenesstas work analysis, that I a trap circuit isconsiderably enhanced.

sym We can now concentrate `on the determination Z ,=M--, (3) fof ztherelationship `betweenthe pole locations (Ym'FYmMYm-IYgohy 40 -and theresonant frequencies of the tuned cirlwhere cuits, asrevidenced by; the.denominator; of(` Equa- Y, is the complex admittance of tuned circuit H,no? 33 settfg this* .denominator equa; t0 Zero YW' isthecomp'lexadmittance of tuned circuit2, "as mdlcated m the' dlscusslon 'ofEquatlon2 We .23,15 thecompiex admittance` of tuned circuwzl Obtaln and'Misaconstant introduced bythe presence 45 019cm i- Mfn 2-{ C21 1C19|C'2tA-i19)"o\-x21 .=,0 oftriodes H, andi T8, andr has no`fu'rther bearing j(9) on thepole andzero analysls to follow' This::is a secondxdegree.equationiinLoneeof i ".t risc-well knonm rthatthe admittance of a,whosefsolutions is tuned circuit,mayibexfurther,dei-ined, .for pur-)(:M (10) l y ,l i r ssion Y P?Se of pole and Zero analysls by the exp ewhichtherefore correspondsto the \p of Equa- V. Y A"Ynf2C'M--7m) "(4)ti'on'2. Thus, one `of the poles will, asithappena Where occur` at theresonant frequency of either .of .'Yanisftheadmittanceoftthetunedcircuitfunder tundrcmts 19 01' 2- l A :consideration :andidentified v,by subscript n, 55 -DlYldmg Equatlon 9 by the'te'm (if-n),We 'Gais-its totalnparallel capacitanceand I fobtam :is itsindividual,complex pole .locationpas here- Cigggogwkw +,C21(Q19+C20) 0,...)(21) :0i 11,)

inbeforeudened. VIn this connection, it must A bekept in mindmhat the.pole location `of each The second solutionl lof the equation willthenbe individual tuned tcircuit, `acting alone, is not 60 1 AMPM,

. wnecessarily ,the rf-same `as that L of the l entire Tijl*- (12)coacting network,4 but Will merely be useful rin (Where solving for thelatter. C p C :substituting this 'form of =Y-Wim appropriate 65 A=Q2l0hgl :(13) "identifying Asubscripts Vfor 4each Y in `Equation` 3 -19 20we obtain The A :roffEquationrl2 :thenscorresponds @to the .z M 4*QCM--MO Y ((54) Assuming `now, sasis'afottentheicase .inprac'tice @tosolutionsv for Equation r9 -andzwith :itxthesznum- .alsumciently l:closerapproximation, i that rcuned 75 ber of poles of the network.

which is located at frequency fps.

Since each A is a complex quantity and since, in any equation, real andimaginary parts must both be equaL'we can determine the frequencycomponents of these )3s by concentrating on their imaginary parts.

or one pole frequency equals the resonantl frequency of tuned circuitI9, as hereinbefore indicated.

Similarly ApZ-may be analyzed to yield Simultaneous examination ofEquations 16 and 17 reveals that it is the high frequency pole It is,then, this latter pole which must, in accordance with the invention, belocated at a frequency between the end of the passband and the zerofrequency. Thus the condition to be met is Vwhere fh is the highfrequency end of the passband corresponding to jtm of Figure 2.

Furthermore AfZl +fl9 A ,F 1 fh (19) Solving inequality (19) for A weobtain A=O21(C19+C20) fh f19 Inequality (20) constitutes the basis forthe design of the amplifier of Figure 4 in accordance with theinvention, since, by conforming with its requirements, the highfrequency pole of the network is placed between the end of the passbandand the zero.

In practice, C19 and C2@ are often constituted by the output capacitanceof triode i1 and the input capacitance of triode I8, respectively, thusyielding values for these components of inequality (20) which are fixedby the choice of tubes. It then remains only to tune circuits IS and 20to the common frequency ,T19 within the passband at which it is desiredto locate the low-frequency pole. Since fn is determined by the desiredbandpass characteristic of the network and far by the frequency of theadjacent channel sound carrier which is to be trapped, the only variablein (20) is C21, which is the total parallel capacitance of tuned circuit2|, as hereinbefore explained. Thus, by selecting the appropriate valuebodiment of the circuit of Figure 4, never reach the zero frequency for,to do so, capacitance C21 would have to be infinite, clearly animpracticable condition. Thus, the requirement of pole locationintermediate the end of the passband Vand the trap frequency is fullymet by satisfying inequality (20) Note, particularly, that not all ofthe Vpoles which characterize the network of Figure 4 occur at theresonant frequency of someone. of the component tuned circuits. It isfor the more general case such as this, where various tuned circuitsinteract to produce a transferV impedance with poles at points otherthan the resonant frequencies, that the characteristics of my novelnetwork must, to be definite and meaningful be defined in terms of itspole and zero pattern.

It will be seen, from the preceding discussion, that several circuitarrangements are available for producing the aforedescribed pole andzero pattern which characterizes my novel coupling network. As stillother embodiments will occur to those skilled inthe art, withoutdeparting from my inventive concept, I desire the latter to be boundedonly by the appended claims.

I claim:

1. An interstage coupling network for transmitting signals within apredetermined frequency band with upper limit at frequency fn, saidnetwork comprising: a pair of input terminals and a pair of outputterminals, a pair of parallelresonant circuits, both tuned to the sainefrequency f1 within said frequency band, one of said pair of circuitsbeing connected between said input terminals and the other of said pairof circuitsvbeing connected between said output terminals, and a thirdparallel-resonant circuit tuned to a frequency f2 higher than fo andconnected between one of said input terminals and one of said outputterminals, the capacitances of said resonant circuits being related bythe expression where C1 is the capacitance of one of said pair ofresonant circuits, C2 is the capacitance of the other of said pair ofresonant circuits and C12 is the capacitance of said third resonantcircuit.

2. In combination: a source of desired and undesired electrical signalsin mutually exclusive frequency bands, said source having a pair ofoutput terminals; and a network for deriving said desired signals fromsaid source and for rejecting said undesired signals, said networkhaving a pair of input terminals connected to said output terminals ofsaid source, said network having a pair of output terminals, and saidnetwork having a transfer impedance between said input and outputterminals characterized by a zero at a frequency within the frequencyband of said undesired signals, by a first pole at a frequency withinthe frequency band of said desired signals and by a second pole betweenthe frequency of said zero and the end frequency of said band of desiredsignals adjacent thereto.

3. In combination: a source of desired electrical signals within arelatively wide frequency band and of undesired electrical signalswithin a relatively narrow frequency band' outside of and near to saidband of desired signals, said source having a pair of output terminals;and a network for deriving said desired signals from said source and forrejecting said undesired signals, said network having a pair of inputterminals connected to said output terminals of said source, saidnetwork having a pair of outputterminals, and said network having atransfer impedance between said input and output terminals characterizedby a zero at a frequency within said frequency band of undesiredsignals, by a first pole at a frequency within said frequency band ofdesired signals'and by a second pole at a frequency between thefrequency of said zero and the end of said frequency band of desiredsignals adjacent to said zero.

4. In combination: a source of desired and undesired electrical signalsin mutually exclusive frequency bands, said source having a pair ofoutput terminals; and a network having a pair of input terminalsconnected to said output terminals of said source and a pair of outputterminals, said network forming a signal path between said network inputterminals and output terminals transmissive of desired signals from saidsource and substantially non-transmissive of undesired signals from saidsource, said network comprising a plurality of impedance elements, eachresonant at a predetermined frequency, one of said impedance elementsbeing connected effectively in series with said signal path and twoothers of said impedance elements being connected effectively in shuntwith said signal path, said series connected element being adjusted toresonance at a frequency Within said band of undesired signals toproduce a zero in the transfer impedance characteristic of said networkat said frequency, and said shunt connected elequency within said bandof desired signals and a f second transfer impedance pole at a frequencybetween said zero frequency and the end of said 14 desired band ofsignals adjacent to said zero frequency.

5. An electrical coupling network forming a signal path transmissive ofsignals in a predeteru mined frequency passband, said network comprisinga plurality of vacuum tubes each having at least triode elements, saidtubes being connested in cascade in said signal path, a first parallelresonant circuit connected in series with said signal path and tuned toresonance at a frequency outside said passband, a second parallelresonant circuit connected in shunt with said signal path and tuned toresonance at a frequency within said passband, and a third parallelresonant circuit connected in shunt with said signal path and tuned toresonance at a frequency between the resonant frequency of said firstcircuit and the end frequency of said passband adjacent thereto, onlyone of said resonant circuits being connected intermediate any two ofsaid cascade-connected vacuum tubes.

References Cited in the le of this patent UNITED STATES PATENTS

